Signal modulation is an operation that changes some of the properties (such as amplitude, frequency, phase, or any combination thereof) of a high frequency sine wave signal, called a carrier signal, as a function of another signal, called a modulating signal. Because the carrier is commonly at a much higher frequency of the modulating signal, this operation is also called up-conversion.
The most common architecture used to modulate a signal at a high frequency (e.g. from a few MHz to tens of GHz) is the Gilbert cell mixer, that acts as a multiplier of two inputs. It essentially multiplies a modulating signal (e.g., a digital signal) applied at a first input with a carrier signal (or a local oscillator signal) applied at a second input, thereby modulating the carrier signal according to the binary data. Usually, the modulating signal is applied at a linear input which performs a voltage-to-current conversion, while the modulated signal—the carrier signal—is applied to a nonlinear input (also called switching input). Driving the mixer's non-linear input with a local oscillator signal (a very high frequency signal) requires the amplitude of the latter to be sufficiently high in order to drive the switching input under any process, voltage, and temperature (PVT) variations, and hence leads to high power consumption. For power efficient modulation, known solutions propose swapping the signals applied at the two inputs of the mixer. Applying the local oscillator signal to the linear input of the mixer allows using a local oscillator signal with a smaller amplitude, thus reducing the power consumption of the mixer.
If the modulating signal is a digital signal (i.e. a binary sequence), the modulated signal is the multiplication of the carrier signal by +1 or −1, depending on the bit sign. This type of modulation is called Binary Phase Shift Keying (BPSK). Without taking any countermeasures, the full spectrum of the digital signal, or all frequency components the digital signal consists of including its respective high-frequency components (its side lobes), are up-converted. However, the side lobes at high frequency may be not compatible with the spectral emission mask requirements, such as the ESTI specification for 79 GHz radar applications, that specify the energy requirements for the frequency components located around an intended operational bandwidth. This issue is explained in more details below.
In perfect BPSK modulation, the high-frequency carrier is multiplied by +1 or −1, depending on the modulating bit. This means the output is a sine wave with a phase of either 0 or 180 degrees (i.e. 0 or π). Such instantaneous phase jumps introduce discontinuities in the waveform of the modulated signal, which are translated in a wide-frequency spectrum with respect to the single tone of the carrier signal, as shown in FIG. 1, thereby causing side lobes to appear around the carrier tone. The first side lobe corresponds to the third harmonic, the second side lobe corresponds to the fifth harmonic, and so forth. The power (P) of the side lobes located around the main lobe (or the intended operational bandwidth) decreases according to:
                                          (                                          sin                ⁡                                  (                  x                  )                                            x                        )                    2                ,                  x          =                      π            ⁢                          f                              f                s                                                                        (        1        )            where f is the offset frequency from the carrier on the x-axis, and fs is the sampling frequency of the baseband signal (2 GHz in FIG. 1).
Since the power of the side lobes (i.e., each side lobe's respective energy content) decreases as shown in (1), the power of a side lobe with respect to the main lobe can be computed in decibels as
                              P          n                =                              -            20                    ⁢                      log            10                    ⁢                      2                          π              ⁡                              (                                                      2                    ⁢                    n                                    +                  1                                )                                                                        (        2        )            where n is the number of the side lobe.
From the above equation, the first and the second side lobes have a power of −13.4 dB and −17.9 dB with respect to the main lobe, respectively, and thus are not compliant with the emission specification set by the ETSI standardization bodies. By way of example, the ETSI emission specification for radar applications in the 79 GHz band allows for side lobes with a power that is 27 dB lower than the power of the main lobe, as shown in FIG. 2.
To comply with the emission mask requirements, conventional solutions opt for either reducing the peak transmitted power by an additional 13.6 dB (27 dB-13.4 dB) or filtering the side lobes.
Some known solutions propose reduction of harmonic content caused by the baseband signal by filtering the signal at the modulator output—filtering at radio frequencies (RF). But achieving a substantial suppression of the side lobes at such high frequencies requires a higher-order band-pass filter with a high quality factor (e.g., higher than the ratio of the required center frequency to required bandwidth). In radar applications where a radio should operates at 80 GHz within a 4 GHz-wide bandwidth, a second order band-pass filter (BPF) with a quality factor of at least 20 (i.e., 80 GHz/4 GHz) is required. Realizing an RF BPF with such high quality factor, good control on the center frequency, and the requisite bandwidth is practically impossible using standard CMOS technology. External filters with lumped or distributed components may be used, but this would unavoidably lead to extra cost and most likely losses. Such solutions are therefore not practical or cost-effective for applications at very high frequencies—frequencies above 5-10 GHz.
Other solutions propose reducing the side lobes by filtering the high-frequency content of the baseband signal prior to up-conversion. A baseband filter smooths the transitions of the baseband waveform from “0” to “1”, leading to a smoother phase transition from 0 to 180 degrees in the waveform at the modulator output. While effective in reducing distant side lobes, baseband filtering is not effective in reducing the most critical side lobes—the ones closest to the main lobe. Higher-order baseband filters may be more effective in reducing the critical side lobes, but such filters require active components (such as transistors, transconductors, operational amplifiers) or inductors to generate complex conjugate poles for a sharper response, and thus leading to higher power consumption and circuit complexity.
Harmonic Rejection (HR) is a technique used for rejecting harmonics caused by the local oscillator signal during the mixing operation. A typical HR technique allows for suppression of the local oscillator's side lobes corresponding to (i) the third harmonic, (ii) the third and fifth harmonics, and/or (iii) the third, fifth and seventh harmonics, depending on the complexity of the applied harmonic rejection. Generally, the higher the harmonic to be suppressed, the more complex the modulator and the local oscillator (or clock) circuit is. HR techniques also require the input signal to be multiplied with copies of the local oscillator signal having different delays (phase) and weights (amplitude).
By properly choosing delays and weights, the summation of the mixer's outputs cancels the harmonics caused by the local oscillator signal. But the HR technique is not applicable for devices operating at millimeter-wave frequencies, where power consumption, area, and circuit complexity are of high importance, as the complexity of the systems employing HR grows exponentially with the number of harmonics to be cancelled.
In paper “A 79 GHz SiGe-Bipolar Spread-Spectrum TX for Automotive Radar” (S. Trotta et al, IEEE Solid-State Circuits Conference, 2007, pp. 430-613, 11-15 Feb. 2007), Trotta et al propose a bi-phase modulator utilizing a Gilbert-cell mixer for short-range automotive radar applications in the 79 GHz band, with the local oscillator signal being applied at the linear (transconductance) input of the mixer. But the architecture suffers from high-side lobes and fails to comply with emission regulations.
In “Sinusoidal SBPSK Modulation Waveform for UHF SATCOM Channels with Improved Adjacent Channel Emissions” (M. A. Belkerdid et al, IEEE Military Communications Conference, pp. 1-7, 29-31 Oct. 2007), Belkerdid et al propose a method for reducing the side-lobes in BPSK modulation by splitting the baseband binary signal to I and Q signals and applying non-linear phase shaping to I and Q signals prior to their modulation. This results in a modulated carrier with a sinusoidal phase transition from 0 to ±180 degrees over a time interval less than the bit period.